FIG. 5 is a circuit diagram of a conventional electric power converter that obtains a DC output insulated from a DC power supply. The conventional electric power converter shown in FIG. 5 has a circuit configuration almost the same with the circuit configurations of the switch mode power supply apparatuses disclosed in Unexamined Patent Application Publication JP 2004-153948 A (U.S. Pat. No. 6,917,528 B2) and Unexamined Patent Application Publication JP 2007-0683597 A (US 2007/0047267 A1). In the circuit shown in FIG. 5, a main switching device (low-side switching device) 1 and a subsidiary switching device (high-side switching device) 2 repeat switching ON and OFF alternately. During the OFF-period of main switching device 1, the circuit shown in FIG. 5 releases the excitation energy stored in an insulation transformer (hereinafter referred to simply as a “transformer”) 6 during the ON-period of main switching device 1 to feed DC electric power to a not-shown load. In FIG. 5, a DC power supply 3 is shown.
Now the operations of the circuit shown in FIG. 5 will be described with reference to FIG. 6 and in connection with metal oxide semiconductor field-effect transistors (hereinafter referred to as “MOSFET's”) employed for main and subsidiary switching devices 1 and 2. In FIG. 6, VGS1, VDS1, and ID1 designate the voltage between the gate and source of main switching device 1, the voltage between the drain and source thereof, and the drain current thereof, respectively. In FIG. 6, VGS2, VDS2, and ID2 designate the voltage between the gate and source of subsidiary switching device 2, the voltage between the drain and source thereof, and the drain current thereof, respectively. In FIG. 6, IDr designates the current of a rectifying diode 8 shown in FIG. 5.
Below, the wave chart in FIG. 6 is divided by time points t1, t2, t3, t4, t40, t5, and t6 into states 1 through 7 and the operations of the conventional electric power converter in state 1 through state 7 between the adjacent time points will be described.
State 1: t1-t2
As the gate input capacitance of main switching device 1 is charged via a resistor 18 with a voltage generated across a third winding 6f and the voltage VGS1 between the gate and source of main switching device 1 exceeds a threshold VGS(th), main switching device 1 is turned ON. The side of third winding 6f opposite to the resistor 18 is connected to the positive side of power supply 3 via resistor 19 and to the negative side of power supply 3 via capacitor 22. In the state in which the body diode of main switching device 1 is conductive such that VDS1 is zero, main switching device 1 conducts zero-voltage turn-ON. The drain current ID1 of main switching device 1, that is equal to the exciting current of transformer 6, increases linearly. Since the voltage VGS2 between the gate and source of subsidiary switching device 2 is negative due to the voltage generated across a fourth winding 6b, subsidiary switching device 2 is in the OFF-state thereof.
State 2: t2-t3
As the voltage generated across a resistor 17 by the drain current ID1 of main switching device 1 exceeds the voltage between the base and emitter of a transistor 21, transistor 21 is turned ON. As transistor 21 is turned ON, the gate input capacitance of main switching device 1 is discharged via a diode 20 and transistor 21, main switching device 1 is turned OFF, VDS1 rises and VDS2 falls. As VDS1 rises, the voltage across fourth winding 6b starts rising.
State 3: t3-t4
Diode 8 becomes conductive and the excitation energy stored in transformer 6 is released to the secondary side. The voltage across fourth winding 6b keeps rising and changes from negative to positive.
State 4: t4-t40
As the voltage across fourth winding 6b exceeds the gate threshold VGS(th) of subsidiary switching device 2 to the higher side, subsidiary switching device 2 conducts zero-voltage turn-ON in the state, in which a current is flowing through the body diode thereof.
State 5: t40-t5
As all the excitation energies stored in transformer 6 are released, diode 8 becomes OFF and the voltage across fourth winding 6b starts falling.
State 6: t5-t6
As the voltage across fourth winding 6b exceeds the gate threshold VGS(th) of subsidiary switching device 2 to the lower side, subsidiary switching device 2 is turned OFF, VDS2 rises, and VDS1 falls.
State 7: t6-
VDS1 becomes zero and VDS2 is the voltage of DC power supply 3. Subsequently, the switch mode power supply returns to the operation of state 1 and repeats the operations in state 1 through state 7, resulting in a self-oscillation.
Since both the main and subsidiary switching devices in the circuit shown in FIG. 6 conduct zero-voltage turn-ON, any turn-ON loss is not caused. Moreover, the excitation energies stored in the transformer leakage inductance and a reactor 5 are regenerated to the DC power supply or to the secondary side during the turn-OFF of any of main and subsidiary switching devices 1 and 2. Therefore, an electric power converter that exhibits low losses and a high conversion efficiency is realized.
However, it is necessary to design the circuit that drives the subsidiary switching device thereof by the auxiliary (fourth) winding of a transformer so that the voltage applied between the gate and source of the subsidiary switching device may not exceed the gate breakdown voltage of the subsidiary switching device. When the subsidiary switching device is a MOSFET, the gate breakdown voltage thereof is around ±30 V generally.
Immediately after DC power supply 3 is fed and main switching device 1 starts switching, the voltage across a capacitor 4 is zero. Therefore, the maximum value VGSmax of the voltage between the gate and source of subsidiary switching device 2 is expressed by the following formula (1).VGSmax=(Voltage of DC power supply 3)×(Number of turns in auxiliary winding 6b)÷(Number of turns in a first winding 6a)  (1)
At some instances other than the start, the voltage as described by the formula (1) is applied between the gate and source of subsidiary switching device 2. When DC power supply 3 is obtained by rectifying a commercial AC power supply, DC power supply 3 is different from country to country, since the AC power supply voltage is different from country to country. If one wants to obtain a switch mode power supply employable under an AC power supply voltage in any country, the range of the voltage across DC power supply 3 will be inevitably wide. For example, the DC power supply voltage in the area, in which the AC power supply voltage is 240 V, is more than twice as high as the DC power supply voltage in the area, in which the AC power supply voltage is 100 V. As the formula (1) clearly indicates, it is very hard to design the number of turns in auxiliary winding 6b so that VGSmax may not exceed the gate breakdown voltage over the entire voltage range of DC power supply 3. For obviating this problem, a Zener diode is connected between the gate and source of the subsidiary switching device to clamp the voltage between the gate and source of the subsidiary switching device with the Zener voltage.
When the voltage between the gate and source of the subsidiary switching device is clamped as described above, a current flows through the Zener diode via a resistor 16 connected to the gate terminal of the subsidiary switching device. As the resistance value of resistor 16 is larger, the gate voltage rises more slowly when the subsidiary switching device is turned ON, causing conduction loss increase in the subsidiary switching device. When the subsidiary switching device is turned OFF, the gate voltage falls slowly, causing turn-OFF loss increase. Therefore, the resistance value of resistor 16 is set from several tens Ω to several hundreds Ω generally. However, when the resistance value of resistor 16 is set from several tens Ω to several hundreds Ω, the problems which impair the reliability of the switch mode power supply are caused. The problems include, for example, a current flowing through the Zener diode that is much higher than the rated current. The problems include also a clamped voltage higher than the Zener voltage described in a data sheet and high enough to exceed the breakdown voltage of the control (gate) terminal of the subsidiary switching device.
For obviating the problems described above, the switch mode power supply disclosed in the JP 2007-0683597 A and described in FIG. 5 attached to the description of the present invention employs a subsidiary control circuit 200a as shown in FIG. 7 and comprising transistor 201, diode 202, Zener diode 203 and resistor 204, to prevent positive and negative over voltages from being applied to the control (gate) terminal of the subsidiary switching device. However, the circuit described in FIG. 7 makes transistor 107 work against a negative over voltage and transistor 201 work against a positive over voltage. Therefore, the circuit described in FIG. 7 employs many constituent parts that add to the manufacturing costs of the switching device.
In view of the foregoing, it would be desirable to obviate the problems described above. It would also desirable to provide a very reliable and inexpensive switch mode power supply that can control the control terminal voltage of a subsidiary switching device to be lower than the gate breakdown voltage over a wide DC input voltage range and in many operation modes.